Ron's Digital Signal Processing Page

Quick & Dirty FIR filter generation

Here's my recipe for a simple Q&D Windowed-Sinc FIR filter generator
in about ten lines of Basic, plus setup and some comments:

		: rem  Some example filter parameters:
fsr = 44100     : rem  set fsr = sample rate
fc  =     0     : rem  set fc = 0 for lowpass
		: rem  set fc = center frequency for bandpass filter
		: rem  set fc = fsr/2 for a highpass
bw  =  3000     : rem  bw = bandwidth, range 0 .. fsr/2 and bw >= fsr/n
		: rem  bw = 3 db corner frequency for a lowpass
		: rem  bw = half the 3 db passband for a bandpass filter
nt  =   128     : rem  nt = number of taps + 1 (nt must be even)
		: rem  nt should be > fsr / bw
		: rem  transition band will be around 3.5 * fsr / nt
		: rem    depending on the window applied and ripple spec.
g   =     1     : rem  g = filter gain for bandpass
		: rem  g = 0.5 , half the gain for a lowpass filter
		: rem  fir() is the result array of FIR taps

dim fir(nt) : fir(0) = 0 : rem fir(1)    is the first tap
			 : rem fir(nt/2) is the middle tap
			 : rem fir(nt-1) is the last tap

rem  R. Nicholson's QDDS FIR filter generator cookbook recipe
rem                 QDDS = Quick, Dirty, Dumb and Short
rem  version 0.6b - 2006-Dec-14, 2007-Sep-30
rem  No warranties implied.  Error checking, optimization, and quality
rem    assessment of the "results" is left as an exercise for the student.
rem  (consider this code Open Source under a BSD style license) 

sub wsfiltgen(fir, nt, fc, bw, g, fsr)
  local i,a,ys,yg,yw,yf				: rem  local variables
  for i = 1 to nt-1
        a  = (i - nt/2) * 2.0*pi*bw/fsr         : rem  scale Sinc width
        ys = 1 : if a <> 0 then ys = sin(a)/a   : rem  calculate Sinc function
        yg = g * (4.0 * bw/fsr)                 : rem  correct window gain
        yw = 0.54 - 0.46 * cos(i * 2.0*pi/nt)   : rem  Hamming window
        yf = cos((i - nt/2) * 2.0*pi*fc/fsr)    : rem  spectral shift to fc
	fir(i) = yf * yw * yg * ys              : rem assign fir coeff.
  next i
end sub
Explanation:

A Sinc function (sin(x)/x) is the Fourier transform of a rectangular function. Using a windowed Sinc is one method of designing a FIR filter, starting with an arbitrary rectangle in the frequency domain, and then creating an approximation of that rectangle's Fourier transform in the time domain by cutting a Sinc's width using a finite width window.

Note that the bw parameter sets the width of a rectangle in the frequency domain. fc sets the location of the center of the rectangle. A Sinc is then the Fourier transform of that frequency domain rectangle into the time domain (or the impulse response). A window is required to make a finite length filter out of a Sinc, which has infinite extent. The filter length nt sets this window width, which inversely affects the filter transition band. As the Sinc gets shortened by the window length, the sides of the rectangle in the frequency domain get less steep, or have a wider transition band. The wider the windowed Sinc, the narrower the transition band.

The type of window function selected (Hamming, von Hann, Blackman, Nuttall, etc.) will affect the pass band and stop band response. A Kaiser window will give you one more degree of freedom, at the cost of computing (or approximating from a table) Bessel functions. Note that the Sinc function itself can also be estimated or approximated from a table for implementation in a system without fast math libraries.

Here's someones java applet which creates similar FIR filters: FIR Digital Filter Design Applet

Quick & Dirty Simple Slow ReSampling technique

rem - QDSS Windowed-Sinc ReSampling subroutine in Basic
rem
rem - This function can also be used for interpolation of FFT results
rem
rem function parameters
rem : x      = new sample point location (relative to old indexes)
rem            (e.g. every other integer for 0.5x decimation)
rem : indat  = original data array
rem : alim   = size of data array
rem : fmax   = low pass filter cutoff frequency
rem : fsr    = sample rate
rem : wnwdth = width of windowed Sinc used as the low pass filter
rem
rem resamp() returns a filtered new sample point

sub resamp(x, indat, alim, fmax, fsr, wnwdth)
  local i,j, r_g,r_w,r_a,r_snc,r_y	: rem some local variables
  r_g = 2 * fmax / fsr           : rem Calc gain correction factor
  r_y = 0
  for i = -wnwdth/2 to (wnwdth/2)-1 : rem For 1 window width
    j       = int(x + i)          : rem Calc input sample index
        : rem calculate von Hann Window. Scale and calculate Sinc
    r_w     = 0.5 - 0.5 * cos(2*pi*(0.5 + (j - x)/wnwdth))
    r_a     = 2*pi*(j - x)*fmax/fsr
    r_snc   = 1  : if (r_a <> 0) then r_snc = sin(r_a)/r_a
    if (j >= 0) and (j < alim) then
      r_y   = r_y + r_g * r_w * r_snc * indat(j)
    endif
  next i
  resamp = r_y                  : rem Return new filtered sample
end sub

rem  - Ron Nicholson's QDSS ReSampler cookbook recipe
rem                 QDSS = Quick, Dirty, Simple and Short
rem  Version 0.1b - 2007-Aug-01
rem  Copyright 2007 Ronald H. Nicholson Jr.
rem  No warranties implied.  Error checking, optimization, and
rem    quality assessment of the "results" is left as an exercise
rem    for the student.
rem  (consider this code Open Source under a BSD style license)
rem  IMHO. YMMV.  http://www.nicholson.com/rhn/dsp.html

Notes:

  • fmax should be less than half of fsr, and less than half of new_fsr (the reciprocal of the x step size).
  • Filter quality increases with a larger window width. The wider the window, the closer fmax can approach half of fsr or new_fsr.
  • Several operations inside the FOR loop can be pre-calculated.
  • There are more optimal windows than the von Hann window.
  • If the x step size is rational the same Window and Sinc values will be recalculated repeatedly. Therefore these values can either be cached, or pre-calculated and stored in a table (polyphase interpolation); or interpolated from a smaller pre-calculated table; or computed from a set of low-order polynomials fitted to each section or lobe between zero-crossings of the windowed Sinc (Farrow). (Performance optimization is left as an exercise for the student).

Bibliography:
CCRMA's Resampling page
CCRMA's Bandlimited Interpolation page

Frequency/Pitch Measurement/Estimation Techniques


Here's a summary of some idea's which I've tried or intend
to try for precisely estimating the pitch or frequencies produced
by musical instruments (e.g for tuning guitars and pianos, etc.).
Many of these are quick and dirty methods without much theoretical
backing, but did seem to work somewhat, at least in a few simple
experiments and test inputs.

Note that musical pitch is a psycho-perceptual phenomna which is
different from peak spectral frequency.  Some very musical sounds
even have a missing fundamental frequency in their audio spectrum.
Some of the techniques in this list other than simple FFT usage
may produce more accurate pitch estimates.

(Unfinished Draft - 2006Mar13,  updated 2009Oct09, 2011Oct02,  rhn)


A list of methods for estimating the frequency of some signal :

 Techniques mostly in the Time Domain  :
 
 0.
 - If there is only one sinusoid with zero noise and measurement error,
    then it might be possible to use 3 samples to solve an equation in
    3 unknowns:  y = A * sin(2*pi * F * t + P)
    for amplitude A, frequency F, reference phase P
    (but note that aliasing can occur depending on the sample spacing)
    Here are derivations of two closed form solutions by C.Turner:
        http://www.claysturner.com/dsp/3pointfrequency.pdf
        http://www.claysturner.com/dsp/4pointfrequency.pdf

 1.
 - Measure the distance between 2 positive zero crossings in samples,
    either by counting samples in a buffer, or by triggering a
    counter/timer to start/stop with a comparator (analog or otherwise).
    The distance between zero crossings could be an estimate of the
    period of a sine wave if there isn't much noise in the signal
    and the count rate is high enough relative to the frequency
    of the signal.  The frequency will be the reciprocal of the period
    distance or time, and the accuracy will be limited by the sample
    spacing.

 2.
 - Linear interpolation between the two samples on either side of 
    zero crossings for a more accurate zero crossing estimate, then
    measure as above between 2 interpolated zero crossings.

 2B
 - Average the distance between zero-crossings over several pairs
    of zero-crossings, perhaps doing least squares fit, and/or
    throwing away outlying measurements first.
    (also see item 14., the Phase Vocoder, since FFT phase is a form
    of regression fit to the zero crossing locations of or near a
    particular frequency content.)

 3.
 - Count positive zero crossings in some sample interval.
    count zero crossing per second to give f +-1 Hz.
    Accuracy depends on the length of the counting interval with
    respect to the period being measured.  
    Band-pass filtering before counting might help remove spurious
    zero-crossings caused by noise.

  Note that noise or interfering signals can change or add additional
  zero crossings, and cause the zero-crossing methods above to produce
  spurious frequency estimation results.  It might be possible to
  remove some of the noise or interference by first filtering the
  signal, e.g. a narrow band bandpass filter centered around the
  frequency of interest.  However, if the Q of this filter is too
  high, one might just end up measuring the frequency at which the
  filter "rings".

 4.  - Autocorrelation
    Find the sample lag with the highest correlation.
    Autocorrelation works better than frequency domain methods
    on signals which are periodic but look nothing like a
    sinusoid (e.g. have a missing fundamental component).
    f = sr / lag

 4B.  - Autocorrelation with resampling for higher lag resolution.
    Resample to a much higher sampling rate, then autocorrelate.
    The set of integer lags will then represent a higher resolution
    in the frequency domain.

 4C.  - Window before or during autocorrelation to reduce edge
    effects.
 
 5.  - Least average squared difference offset (ASDF lag)
    Find the sample lag with the lowest difference.
    Works with signals that have near constant amplitude.
    Sometimes normalizing the amplitude (AGC) before calculating
    the ASDF lags helps.
    AMDF - or computing the mean absolute value difference of
    a range of lags is another variation on this approach.

 5B.  - ASDF minima lag with parabolic interpolation
    Interpolate between the three best neighboring lags.
    Parabolic interpolation works best if the "continuous" lag
    curve matches the interpolation kernel, but it's not an
    exact fit using a parabola.

 21.  - Hilbert transform to the complex time domain 
    Use a Hilbert transform filter process to produce an
    imaginary time domain component.  Measure the rate of
    change of phase of the resulting complex vector (real and
    imaginary) to obtain instantaneous frequency.  May need
    to window and bandpass filter some length of signal before
    the Hilbert filter and low-pass filter the resulting phase
    to produce reasonable results.  This technique seems to
    be related to quadrature demodulation at baseband.

 Techniques mostly in the Frequency Domain (using a DFT/FFT) :

 6.  - FFT magnitude peak
    Take the magnitude of the complex FFT = sqrt(b_r^2 +
    b_i^2), Search for the maxima.  i  = index of FFT bin with
    maximum (abs) magnitude peak sr = sample rate (samples
    per second) n  = number of FFT points f = i * ( sr / n )

 7.
 - Window before FFT to reduce the side-lobes of interference:
    // von Hann
    w[i] = 0.5  - 0.5  * cos( 2*PI * i / (n-1) )
    // Hamming
    w[i] = 0.54 - 0.46 * cos( 2*pi * i / (n-1) )
    // Blackman-Nuttall
    w[i] =
        0.3635819
      - 0.4891775*cos(2*pi*i/(n-1))
      + 0.1365995*cos(4*pi*i/(n-1))
      - 0.0106411*cos(6*pi*i/(n-1))

    The FFT of a non-bin frequency sinusiod (for a frequency
    Note that windowing removes information about the signal,
    especially near the sides, and makes the main lobe wider
    and less sharp.  So windowing trades off lower sinusoid
    frequency resolution versus reducing interference from
    nearby signals and/or noise.

 Sub-bin resolution FFT/DFT frequency domain methods

    The FFT of a non-bin frequency sinusiod (for a frequency
    not exactly periodic in the FFT aperture), instead of appearing
    in one bin, will result in a sampled Sinc, or a sampled transform
    of the applied window for non-rectangular windows.  Estimating
    the shift of this resulting Sinc (or other) shape may produce
    a frequency resolution much finer than FFT bin resolution.
 
 8A.
 - Parabolic interpolation of FFT magnitude peak
    where m2 = magnitude of bin nearest peak m1,m3 = magnitude of bins
    to the left and right:
    d = 0.5 * (m1 - m3) / (m1 - 2.0 * m2 + m3)
    f = ( i + d ) * ( sr / n ) 
    Works best using a window that has a parabolic shaped main lobe
    in the frequency domain (Hamming or von Hann window might be close
    enough).
    See:
    http://ccrma.stanford.edu/~jos/parshl/Peak_Detection_Steps_3.html
    for derivation of this interpolation formula.

 8B.
 - Triangle interpolation of FFT magnitude peak
    using a window which has a triangle shaped main lobe
    in the frequency domain so that the FFT peak is sharper.

 9.
 - Sinc interpolation of un-windowed FFT
    Find the best fitting Sinc function to the complex FFT results
    by autocorrelation or least-squares successive approximation.
    Sinc = sin(2*pi*b)/(2*pi*b)
    fc = b * ( sr / n )
    This method allows one to use more bins (try 5 bins containing
    the peak) for interpolation than the parabolic method (only 3
    bins) on the complex FFT result vector. More bins should contains
    more information about a signals frequency, especially if the FFT
    is done without a (non-rectangular) window to reduce side-lobes.
    The Sinc is the Fourier transform of a rectangle.  The rectangle
    results from the finite aperture window of a DFT/FFT chopping
    a signal whether or not it is not periodic in that aperture.
    Variation:
    Estimate the Sinc fit separately on the real and imaginary
    component vectors of the FFT, and then take a weighted average.

 Changing the size of the FFT :
 
 10.
 - Zero-padded FFT 
    use a 32x longer FFT vector with 31x zeros appended
    (or other amount of padding).
    This is mostly a very accurate way to interpolate an
    FFT magnitude peak (a form of Sinc interpolation,
    but modifying in the time domain).
    Works well with a window and additional parabolic interpolation,
    or with no window and Sinc interpolation (the needed Sinc gets
    wider by the vector expansion factor).
    Note that this might increase the accuracy of finding a single
    sinusoid, but doesn't help in the separation of very close
    sinusoids or adjacent spectral interference or noise.
    (Note #2: Zero-padded FFTs are most accurate near Fs/4.
    At very low frequencies, there can be interference from the
    negative frequency complex conjugate image.  2007-Feb-17)

 Using more than one set of FFT results from the same data stream :
 
 11.
 - FFT of windowed derivative
    dx = w(i) * (x[i] - x[i-1])
    f = (sr/pi) * asin(( abs(max_fft_dx)/abs(max_fft_x) ) / 2)
    (seems to work better than Phase Vocoder for very low frequencies)

 12.
 - time-frequency reassignment
    Using multiple windowed FFT's to reassign the center of bins.
    (less popular because the results aren't exactly invertible).
    FFT with the Hann window & find the peak bin = i_fft 
    fc = i_fft * sr / n 
    Calculate fft2 using the derivative of a Hann window
    Calculate fft3 using the Hann window * a ramp from (-1 to 1)
    (see Jez Wells' paper for descriptions of these windows:
    http://www-users.york.ac.uk/~jjw100/ReadingTheSines.pdf )

    f = fc + (sr/(2*pi)) * im( i_fft c* i_fft2 ) / mag( i_fft )
    t = re( i_fft c* i_fft3 ) / mag( i_fft )
    t is the centroid of the sinusoid's amplitude "hump"

 13.
 - Resampling (by windowed Sinc interpolation) to produce a signal
    with a period which is an exact sub-multiple of the FFT size.
    (e.g. successive approximation to center a spectral peak into
    a single bin)
 
 Frequency domanin analysis using more than 1 FFT frame on the data.
 
 14.
 - 2 FFT's + Phase Vocoder analysis for sequential or overlapping frames.
    This method uses phase information between two FFT's with different
    offset (disjoint or overlapping) apertures into the data, and
    assuming that the same peak bin appears in both FFT frames.

    i = nearest FFT bin# to peak, (initial f estimate = (i * sr / n))
    ph0 = atan2( imag(i), real(i) )
    ph1 = phase in second FFT frame or aperture
    p = ( ph0 + i * 2 * pi * offset / n ) - ph1
    Unwrap the phase offset p to between -pi and pi
    The phase vocoder corrected frequency estimate is then:
    f = (i * sr / n) - (p * sr / (2 * pi * offset))

    Try a von Hann or Blackman-Nuttall window for better accuracy.
    (equation corrected 2008Jun19 by rhn)

    Note that a phase vocoder can be considered a fancy method of
    measuring zero crossings, but using the phase of the correlation
    against a sinusoid to get a better statistical estimate of where
    multiple zero crossings of the frequency should have been located.
    Note that you can interpolate phases between FFT bins by using
    something like FFTShift to (re)reference the phases (real = even,
    imag = odd) to the center of the FFT aperture.
    (Note #2: Phase Vocoder accuracy is highest if the frame offset
    is close to a multiple of a full period of the frequency being
    estimated.  This is because the shape of the "spectral leakage"
    is slightly different for odd/even sinusoids, especially for
    relatively low frequency sinusoids. 2008-Feb-18 )

 14B
 - FFT + an 1 offset dft bin for phase vocoder
    phase vocoder uses successive FFT's
    If only one frequency is needed then only one 
    DFT bin (complex correlation) needs to be calculated.

 14C
 - FFT + using Goertzel algorithm for dft bin for phase vocoder
    This is similar to an optimization for doing a 1-bin DFT.
    Goertzel may require > 32 bits precision for low f0/sr ratios.

 Techniques mostly in the Quefrency Domain  :

 16
 - Cepstrum
     Look for excitation peaks in ifft(log(mag(fft(samples))))
     Useful for estimating the pitch of signals rich
     in harmonics, but possibly with a weak or missing
     fundamental component.
	see: http://cnx.org/content/m12469/latest/
     Used in speech and singing research.

   If a signal has a train of harmonics, then one can try averaging
   the estimated frequency of several harmonics divided by their
   harmonic number (try several octave number offsets).  This is a
   crude variation on using Cepstral peaks.

 16B
 - Harmonic Product Spectrum
     Compress a spectrum by 2X, 3X, 4X, etc. and overlay the spectra.
     A signal producing those harmonics will produce overlapping
     peaks in the overlay, with the maximum number of peaks
     overlapping at what could be a fundamental frequency.
     The low frequency spectra or the fundamental may need to be
     interpolated.  Also consider the case for missing fundamentals.
     see:
     http://musicweb.ucsd.edu/~trsmyth/analysis/Harmonic_Product_Spectrum.html

 Other Methods:
 
 15
 - continuous quadrature (de)modulation with phase extraction
   (or a digital frequency discriminator)
    Fit a slope to the phase data to produce a frequency estimate.
    (frequency is the derivative of phase). Perhaps use
    least squares and/or low-degree spline fit to filter out noise.
    Could use a table lookup or Goertzel for the modulating signal.
    This seems like a boundary case of a 1-sample phase vocoding,
    using quadrature multiplication instead of an FFT to extract
    phase information.
    This is how frequency is estimated in some FM radio and modem
    demodulation schemes.

 15B
 - PLL
   Try locking a digitally synthesized PLL oscillator to the data.
   If it locks, use the synthesized frequency as the estimate.
   For stored or buffered data, the PLL can also be run backwards
   in "time", or even in both directions for constant frequency
   sources.

Other methods (which I haven't tried, YMMV):

 21 
 - RAPT (Robust Algorithm for Pitch Tracking)
    Appears to be similar to Harmonic Product Spectrum, except that
    it uses cross-correlation lags at multiple resolutions instead
    of FFT spectra, plus dynamic programming to minimize a weighting
    function over several analysis frames.
    see:
    https://www.ee.columbia.edu/~dpwe/papers/Talkin95-rapt.pdf
    
    and:
    http://dsp.tut.su/irapt.html

 21B
 - YAAPT (Yet Another Algorithm for Pitch Tracking)
    Uses Normalized Cross Correlation (NCCF), peak picking, and
    dynamic programming using transition costs.
    see:
    http://bingweb.binghamton.edu/~hhu1/pitch/YAPT.pdf

 17
 - MUSIC
 	see: http://cnx.org/content/m10588/latest/

 18
 - ESPIRIT
 	another matrix eigenvector decomposition approach

 19
 - Exponential decaying phasor curve fitting

 20
 - State Space embedding
    Useful if you are looking at a mildly bifurcated chaotic type
    of signal.
  
 References: 
 See also:
 	MIREX / Music Information Retrieval - Melody Extraction:
	http://www.music-ir.org/mirex/wiki/2014:Audio_Melody_Extraction
	Klapuri's publications on polyphonic pitch extraction:
	http://www.cs.tut.fi/~klap/iiro/
	Wikipedia page on Pitch Detection/Estimation:
	https://en.wikipedia.org/wiki/Pitch_detection_algorithm

Bandwidth of a Goertzel filter (or 1 zero-padded DFT bin)

The Goertzel algorithm produces about the same thing as the magnitude from one bin of a DFT/FFT (there will be differences in numerical accuracy). The bandwidth of one DFT bin will vary inversely proportionally to the length N of the DFT/FFT (with the bin spacing being fs/N).

A DFT, or Goertzel filter, for a finite length data vector, acts as if a rectangular window was applied to the data. Thus the resulting spectrum is roughly a Sinc function (sin(x)/x), with a main lobe plus symmetric decaying ripples to each side.

But, more precisely, the result for real-valued data is the sum of two Dirichlet kernels, or periodic Sinc's. This is because: (1) the spectrum of sampled data repeats in the frequency domain; and (2) the spectrum of real data includes two mirrored complex conjugate spectrums. So you end up with two periodic trains of Sinc functions, repeating at the frequency of the window period, each periodic train of the opposite phase to the other.

So, including these effects for the one bin of a DFT, zero-padded DFT, or a Goertzel filter, I get this more precise result for the frequency response: 

Given:
  ft = frequency of test sinusoid
  sr = sample rate
  n  is the length of the rectangular window (left justified)

  Main lobe magnitude at frequency f:
  Hm(f) = sin(n * pi*(ft-f)/sr) / (n * sin(pi*(ft-f)/sr))
  (this is a periodic Sinc, or Dirichlet kernel)

  Complex conjugate lobe magnitude at f:
  Hc(f) = sin(n * pi*(ft+f)/sr) / (n * sin(pi*(ft+f)/sr))

  The DFT/FFT magnitude sum:
  M(f) = Hm(f)^2 + Hc(f)^2 + 2*Hm(f)*Hc(f)*cos(2 * ph(ft))

Where ph(ft) is the phase of the sinusoid relative to the center of rectangular window (it should be 0 for sinusoid that is an even function in the rectangular window.)

The correction factor Hc is "small" if the bin number (f0/sr) is "big", and thus usually ignored.

Also note that n * sin(w) is close to n * w for very small w, thus allowing simplifying the equations to the standard Sinc formulation of H(f) = sin(n*w)/(n*w).

Note that the phase of the data, relative to the window center, does make a difference for spectra that is not strictly periodic in the window or aperture. This is due to fact that the negative frequency Sinc convolution is the complex conjugate of the positive frequency Sinc convolution for real data, thus the two types of Sinc (real & imaginary) sum oppositely.

This equation seems to work even if neither f or f0 is periodic in a window of length n. That turns out to be another difference between a 1-bin DFT and Goertzel. A DFT/FFT only calculates correlations against periodic sinusoids in the DFT aperature. A Goertzel filter can be used for correlation against any frequency, irrespective of window length or periodicity.

The above equation can also be used to compute the magnitude spectrum for the FFT of zero-padded data (with the padding added after the data, or on the right), or equivalently rectangularly windowed data, by setting n in the above equation to be the width of the rectangular data window instead of the width of the whole FFT. The even/oddness phase should be measured or estimated relative to the center of the rectangular window, not the DFT aperture.

(revised 2008-Feb-19, rhn)

Some Common FFT Misconceptions


Just for fun, I thought I'd try and list many of the
misconceptions that people seem to have regarding using
the FFT, as seen in various previous posts to comp.dsp.

The FFT is a widely used algorithm because the result has
something (not precisely understood) to do with frequencies.

Some Common FFT Misconceptions:

1.
The FFT will directly produce the frequency of some
phenomena, without regard to the relationship between FFT
length versus the period of any oscillations (e.g. what
rectangular window?), or any envelope on or modulation
of some periodic waveform.

2.
The FFT will find the correct frequency content without
regard to the relationship between the sample rate and the
highest frequency content present in the sampled phenomena,
or any other need for bandlimiting (for example, of closing
price data).
2b.
The sample rate only needs to be twice the highest
present, or twice the highest  frequency of interest for an
FFT to "find" that frequency.

3.
Since the FFT always produces "frequencies", there must
be some periodic phenomena within the sampled data.

4.
The perceive pitch of a musical or speech waveform is the
same as the frequency represented by the FFT magnitude peak.

5.
Windowing (non-rectangular) will always result in a more
accurate frequency estimation.

6.
One can perfectly filter out a signal band by just zeroing
out some bins between an FFT and IFFT.

7.
One can filter in the frequency domain without regard to the
length of the (significant) impulse response of the filter.

8.
One can reconstruct a signal from just the FFT magnitude
result.

9.
One can reconstruct a real signal by just feeding the
"positive frequency" bins to a generic IFFT.

10.
One can extrapolate a signal from an FFT without inferring
circular boundary conditions.

11.
The (complex) FFT contains no information about the time
domain envelope.

11.
The phase of a sinusoid (or how far that sinusoid is away
from Fs/4) does not affect the magnitude result of an FFT.

12.
Information from preceding, overlapping, or different length
FFT frames is of no interest in analyzing the spectrum of
some waveform.

13.
The "resolution" of a frequency estimation depends only
on the FFT length, and has nothing to do with anything
known about the signal-to-noise ratio in the sampled
data.
13b.
More "resolution" of frequency estimation is created by
zero-padding (...beyond that provided by other methods
of interpolation).

14.
An FFT is not just an efficient implementation of a DFT.
(Or: "What's a DFT?")

15.
Using an entire FFT is the best way to check for even a
very small number of frequencies (say a DTMF tone).

16.
The FFT can only operate on vector lengths that are an
integer power of 2.  (Or FFT's of vectors not a power of
2 are "really" slow.)

17.
The performance of an FFT implementation (on contemporary
PC's) depends mostly on the number of arithmetic operations
(or MACs or FMACs).

18.
An FFT is less accurate than using the defining DFT
formula for computation.

19.
???

The above are common FFT misconceptions, or sometimes just
poorly thought out or stated conceptions that can lead to
strange misinterpretations of the results of using an FFT.

First posted: 2008-March-28 to comp.dsp
Last edited: 2008-March-31

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& email: rhn@nicholson.com
I also post in comp.dsp signing as: rhn A.T nicholson d.0.t C-o-M
Copyright 2006,2008,2011,2012 Ronald H Nicholson, Jr.